AP6503340kHz 23V 3A SYNCHRONOUS DC/DC BUCK CONVERTER
Description
The AP6503 is a 340kHz switching frequency external compensated synchronous DC/DC buck converter. It has integrated low RDSON high and low side MOSFETs.
Pin Assignments
( Top View )
BSINSWGND
12348765SSENCOMPFB
The AP6503 enables continues load current of up to 3A with efficiency as high as 95%.
The AP6503 features current mode control operation, which enables fast transient response times and easy loop stabilization.
The AP6503 simplifies board layout and reduces space requirements with its high level of integration and minimal need for external components, making it ideal for distributed power architectures.
The AP6503 is available in a standard Green SO-8EP package with exposed PAD for improved thermal performance and is RoHS compliant.
SO-8EP
Features
• • • •
VIN 4.7V to 23V
3A continuous Output Current, 4A Peak VOUT adjustable to 0.925 to 20V 340kHz switching frequency
Applications
• Gaming Consoles • • •
Flat Screen TV sets and Monitors Set Top Boxes
Distributed power systems
• Programmable Soft-Start • Enable pin • Protection
OCP
Thermal Shutdown • •
Notes:
• Home Audio
• Consumer electronics • Network Systems •
FPGA, DSP and ASIC Supplies • Green Electronics
Totally Lead-Free & Fully RoHS Compliant (Notes 1 & 2) Halogen and Antimony Free. “Green” Device (Note 3)
1. No purposely added lead. Fully EU Directive 2002/95/EC (RoHS) & 2011/65/EU (RoHS 2) compliant.
2. See http://www.diodes.com for more information about Diodes Incorporated’s definitions of Halogen- and Antimony-free, \"Green\" and Lead-free.
3. Halogen- and Antimony-free \"Green” products are defined as those which contain <900ppm bromine, <900ppm chlorine (<1500ppm total Br + Cl) and <1000ppm antimony compounds.
Typical Applications Circuit
10090VIN =5VVIN = 12VEFFICIENCY (%)80706050VOUT = 3.3VL = 10µH40012LOAD CURRENT (A)Efficiency vs. Load Current3Figure 1 Typical Application Circuit
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AP6503
Document number: DS35077 Rev. 5 - 2
AP6503
Pin Descriptions
Pin Number
Pin Name
Function
High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET switch.
1 BS
Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch.
Power Input. IN supplies the power to the IC, as well as the step-down converter switches. Drive IN with a
2 IN 4.7V to 23V power source. Bypass IN to GND with a suitably large capacitor to eliminate noise on the input
to the IC. See Input Capacitor.
Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC
3 SW filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side
switch.c
4 GND Ground (Connect the exposed pad to Pin 4).
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage divider
5 FB
connected to it from the output voltage. The feedback threshold is 0.925V. See Setting the Output Voltage. Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC
6 COMP network from COMP to GND. In some cases, an additional capacitor from COMP to GND is required. See
Compensation Components.
Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on the regulator;
7 EN
low to turn it off. Attach to IN with a 100kΩ pull up resistor for automatic startup.
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the
8 SS soft-start period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the soft-start feature, leave
SS floating.
EP EP EP exposed thermal pad connect to Pin 4 GND
Functional Block Diagram
+OVP-1.1VOSCILLATOR+100/340 KHz-0.3 V+-SS80.923 V++ERROR AMPLIFIER6uA-CURRENTCOMPARATORCLKLogic+INEN7+INTERNALREGULATORS5V-1100mΩ3100mΩSWBS4COMP62.5V+EN OKLOCKOUTCOMPARATORdisableIN < 4.10VGNDRAMPECURRENT SENSEAMPLIFIER2INFB5-0.9V-SHUTDOWN COMPARATOR
Figure 2 Functional Block Diagram
AP6503
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Absolute Maximum Ratings (Note 4) (@TA = +25°C, unless otherwise specified.)
Symbol Parameter VIN VSW VBS VFB VEN VCOMP TST TJ TL HBM
Notes:
Rating Unit -0.3 to +26 -1.0 to VIN +0.3 VSW -0.3 to VSW +6
-0.3V to +6 -0.3V to +6 -0.3V to +6 -65 to +150 +150 +260
V V V V V V °C °C °C
Supply Voltage Switch Node Voltage Bootstrap Voltage Feedback Voltage Enable/UVLO Voltage Comp Voltage Storage Temperature Junction Temperature Lead Temperature Human Body Model
ESD Susceptibility (Note 5)
MM Machine Model
250
V 3 kV 4. Stresses greater than the 'Absolute Maximum Ratings' specified above, may cause permanent damage to the device. These are stress ratings only;
functional operation of the device at these or any other conditions exceeding those indicated in this specification is not implied. Device reliability may be affected by exposure to absolute maximum rating conditions for extended periods of time.
5. Semiconductor devices are ESD sensitive and may be damaged by exposure to ESD events. Suitable ESD precautions should be taken when handling and transporting these devices.
Thermal Resistance (Note 6)
Symbol Parameter θJA θJC
Note:
Rating Unit 74 16
°C/W °C/W
Junction to Ambient Junction to Case
6. Test condition for SO-8EP: Measured on approximately 1” square of 1 oz copper
Recommended Operating Conditions (Note 7) (@TA = +25°C, unless otherwise specified.)
VIN TA
Note:
Symbol Parameter Min Max Unit Supply Voltage
Operating Ambient Temperature Range
4.7 -40
23 +85
V °C
7. The device function is not guaranteed outside of the recommended operating conditions.
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AP6503
Electrical Characteristics (VIN = 12V, TA = +25°C, unless otherwise specified.)
Symbol Parameter Test Conditions Min Typ Max Unit Shutdown Supply Current 0.3 3.0 µA IIN VEN = 0V Supply Current (Quiescent) 0.6 1.5 mA IIN VEN = 2.0V, VFB = 1.0V High-Side Switch On-Resistance (Note 8) 100 mΩ RDS(ON)1
Low-Side Switch On-Resistance (Note 8) 100 mΩ RDS(ON)2
HS Current Limit Minimum duty cycle 5.5 A ILimit
LS Current Limit From Drain to Source 0.9 A ILimit
VEN = 0V, VSW = 0V,
0 10 μA High-Side Switch Leakage Current
VSW =12V
Error Amplifier Voltage Gain
AVEA 800 V/V (Note 8)
GEA Error Amplifier Transconductance 1000 µA/V ΔIC = ±10µA
COMP to Current Sense
GCS 2.8 A/V Transconductance Oscillator Frequency 300 340 380 kHz FSW VFB = 0.75V Fold-back Frequency 0.30 FFB VFB = 0V fSW Maximum Duty Cycle 90 % DMAX VFB = 800mV Minimum On Time 130 ns TON
Feedback Voltage 900 925 950 mV VFB TA = -40°C to +85°C
Feedback Overvoltage Threshold 1.1 V
EN Rising Threshold 0.7 0.8 0.9 V VEN_Rising
EN Lockout Threshold Voltage 2.2 2.5 2.7 V EN Lockout Hysteresis 220 mV
3.80 4.05 4.40 V INUVVth VIN Under Voltage Threshold Rising
250 mV INUVHYS VIN Under Voltage Threshold Hysteresis
Soft-Start Current 6 μA VSS = 0V Soft-Start Period 15 ms CSS = 0.1µF
Thermal Shutdown (Note 8) 160 °C TSD
Note: 8. Guaranteed by design
AP6503
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Typical Performance Graphs (VIN = 12V, VOUT = 3.3V, TA = +25°C, unless otherwise noted)
0.6QUIESCENT SUPPLY CURRENT (mA)0.074SHUTDOWN SUPPLY CURRENT (µA)05101520250.580.560.00.0540.0440.0340.0240.0140.0040510152025INPUT VOLTAGE (V)Shutdown Supply Current vs. Input Voltage0.540.520.50.48INPUT VOLTAGE (V)Quiescent Supply Current vs. Input Voltage
3.333.3293.328OUTPUT VOLTAGE (V)
6.265.85.65.45.254.8-603.3273.3263.3253.3243.3233.3223.3213.324.759.7514.7519.75INPUT VOLTAGE (V)Line Regulation24.75VIN =12V-40020406080TEMPERATURE (C)Current Limit vs. Temperature-20100
375OSCILLATOR FREQUENCY (Khz)
0.920.9180.916FEEDBACK VOLTAGE (V)0.9140.9120.910.9080.9060.9040.9020.9-60370365360355350-40020406080TEMPERATURE (°C)Feedback Voltage vs. Temperature-20100TEMPERATURE (°C)Oscillator Frequency vs. Temperature
AP6503
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Typical Performance Graphs (cont.) (VIN = 12V, VOUT = 3.3V, TA = +25°C, unless otherwise noted)
9085807570656055504540LOAD CURRENT (A)Efficiency vs. Load CurrentVOUT = 1.2VL = 3.3µHVIN = 5V90858075EFFICIENCY (%)70656055504540LOAD CURRENT (A)Efficiency vs. Load CurrentVOUT = 1.8VL = 3.3µHVIN = 5VVIN = 12VEFFICIENCY (%)VIN = 12V
100
9080EFFICIENCY (%)7060VIN = 12VVOUT =5VL = 10µH50400.20.611.41.82.22.63LOAD CURRENT (A)Efficiency vs. Load Current
AP6503
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Typical Performance Characteristics
(VIN = 12V, VOUT = 3.3V ,L = 3.3µH, C1 = 22µF, C2 = 47µF, TA = +25°C, unless otherwise noted)
Steady State Test no load
Steady State Test 3A
Startup Through Enable_no load
Time -2µs/div
Time -2µs/div
Time -10ms/div
Startup Through Enable 3A Shutdown Through Enable_no loadShutdown Through Enable3A
Time -10ms/div
Load Transient Test 1.5 to 3A
Time -10ms/div Short Circuit Test
Time -5ms/div Short Circuit Recovery
Time -100µs/div
Time -20µs/div Time -20µs/div
AP6503
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AP6503
Application Information
Theory of Operation
The AP6503 is a 3A current mode control, synchronous buck regulator with built in power MOSFETs. Current mode control assures excellent line and load regulation and a wide loop bandwidth for fast response to load transients. Figure 3 depicts the functional block diagram of AP6503. The operation of one switching cycle can be explained as follows. At the beginning of each cycle, HS (high-side) MOSFET is off. The EA output voltage is higher than the current sense amplifier output, and the current comparator’s output is low. The rising edge of the 340kHz oscillator clock signal sets the RS Flip-Flop. Its output turns on HS MOSFET. The current sense amplifier is reset for every switching cycle.
When the HS MOSFET is on, inductor current starts to increase. The Current Sense Amplifier senses and amplifies the inductor current. Since the current mode control is subject to sub-harmonic oscillations that peak at half the switching frequency, Ramp slope compensation is utilized. This will help to stabilize the power supply. This Ramp compensation is summed to the Current Sense Amplifier output and compared to the Error Amplifier output by the PWM Comparator. When the sum of the Current Sense Amplifier output and the Slope Compensation signal exceeds the EA output voltage, the RS Flip-Flop is reset and HS MOSFET is turned off.
For one whole cycle, if the sum of the Current Sense Amplifier output and the Slope Compensation signal does not exceed the EA output, then the falling edge of the oscillator clock resets the Flip-Flop. The output of the Error Amplifier increases when feedback voltage (VFB) is lower than the reference voltage of 0.925V. This also increases the inductor current as it is proportional to the EA voltage.
If in one cycle the current in the power MOSFET does not reach the COMP set current value, the power MOSFET will be forced to turn off. When the HS MOSFET turns off, the synchronous LS MOSFET turns on until the next clock cycle begins. There is a “dead time” between the HS turn off and LS turn on that prevents the switches from “shooting through” from the input supply to ground.
The voltage loop is compensated through an internal transconductance amplifier and can be adjusted through the external compensation components.
Enable
Above the ‘EN Rising Threshold’, the internal regulator is turned on and the quiescent current can be measured above this threshold. The enable (EN) input allows the user to control turning on or off the regulator. To enable the AP6503, EN must be pulled above the ‘EN Lockout Threshold Voltage’ and to disable the AP6503, EN must be pulled below ‘EN Lockout Threshold Voltage - EN Lockout Hysteresis’ (2.2V-0.22V =1.98V).
External Soft Start
Soft start is traditionally implemented to prevent the excess inrush current. This in turn prevents the converter output voltage from overshooting when it reaches regulation. The AP6503 has an internal current source with a soft start capacitor to ramp the reference voltage from 0V to 0.925V. The soft start current is 6uA. The soft start sequence is reset when there is a Thermal Shutdown, Under Voltage Lockout (UVLO) or when the part is disabled using the EN pin.
External Soft Start can be calculated from the formula below:
ISS=C*
Where;
ISS = Soft Start Current C = External Capacitor
DV=change in feedback voltage from 0V to maximum voltage DT = Soft Start Time
DV
DT
Current Limit Protection
In order to reduce the total power dissipation and to protect the application, AP6503 has cycle-by-cycle current limiting implementation. The voltage drop across the internal high-side MOSFET is sensed and compared with the internally set current limit threshold. This voltage drop is sensed at about 30ns after the HS turns on. When the peak inductor current exceeds the set current limit threshold, current limit protection is activated. During this time the feedback voltage (VFB) drops down. When the voltage at the FB pin reaches 0.3V, the internal oscillator shifts the frequency from the normal operating frequency of 340Khz to a fold-back frequency of 102Khz. The current limit is reduced to 70% of nominal current limit when the part is operating at 102Khz. This low Fold-back frequency prevents runaway current.
Under Voltage Lockout (UVLO)
Under Voltage Lockout is implemented to prevent the IC from insufficient input voltages. The AP6503 has a UVLO comparator that monitors the input voltage and the internal bandgap reference. If the input voltage falls below 4.0V, the AP6503 will latch an under voltage fault. In this event the output will be pulled low and power has to be re-cycled to reset the UVLO fault.
AP6503
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AP6503
Application Information (cont.)
Over Voltage Protection
When the AP6503 FB pin exceeds 20% of the nominal regulation voltage of 0.925V, the over voltage comparator is tripped and the COMP pin and the SS pin are discharged to GND, forcing the high-side switch off.
Thermal Shutdown
The AP6503 has on-chip thermal protection that prevents damage to the IC when the die temperature exceeds safe margins. It implements a thermal sensing to monitor the operating junction temperature of the IC. Once the die temperature rises to approximately 160°C, the thermal protection feature gets activated. The internal thermal sense circuitry turns the IC off thus preventing the power switch from damage.
A hysteresis in the thermal sense circuit allows the device to cool down to approximately +120°C before the IC is enabled again through soft start. This thermal hysteresis feature prevents undesirable oscillations of the thermal protection circuit.
Setting the Output Voltage
The output voltage can be adjusted from 0.925V to 18V using an external resistor divider. Table 1 shows a list of resistor selection for common output voltages. Resistor R1 is selected based on a design tradeoff between efficiency and output voltage accuracy. For high values of R1 there is less current consumption in the feedback network. However the trade off is output voltage accuracy due to the bias current in the error amplifier. R1 can be determined by the following equation:
⎛VOUT⎞
R1=R2⋅⎜⎜0.925−1⎟⎟
⎝⎠ Figure 3. Feedback Divider Network
When output voltage is low, network as shown in Figure 3 is recommended.
VOUT(V)
R1(KΩ) R2(KΩ)
5 45.3 10 3.3 26.1 10 2.5 16.9 10 1.8 9.53 10 1.2 3 10
Table 1 – Resistor Selection for Common Output Voltages
Compensation Components
The AP6503 has an external COMP pin through which system stability and transient response can be controlled. COMP pin is the output of the internal trans-conductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC gain of the voltage feedback loop is given by:
AVDC=RLOAD×GCS×AVEA×
VFB
VOUT
Where VFB is the feedback voltage (0.925V), RLOAD is the load resistor value, GCS is the current sense trans-conductance and AVEA is the error amplifier voltage gain.
AP6503
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Application Information (cont.)
Compensation Components (cont.)
The control loop transfer function incorporates two poles one is due to the compensation capacitor (C3) and the output resistor of error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at:
fP1=fP2=
GEA
2π×C3×AVEA1
2π×C2×RLOAD
Where GEA is the error amplifier trans-conductance.
One zero is present due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:
fZ1=
1
2π×C3×R3
The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is crucial.
A rule of thumb is to set the crossover frequency to below one-tenth of the switching frequency. Use the following procedure to optimize the compensation components:
1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation:
R3=
V2π×C2×fc2π×C2×0.1×fsVOUT
××OUT<
GEA×GCSVFBG×GVFB
CSEA
Where fC is the crossover frequency, which is typically less than one tenth of the switching frequency.
2. Choose the compensation capacitor (C3) to achieve the desired phase margin set the compensation zero, fZ1, to below one forth of the crossover frequency to provide sufficient phase margin. Determine the C3 value by the following equation:
C3>
Where R3 is the compensation resistor value.
VOUT (V)
CIN/C1 (µF)
2
π×R3×fc
COUT/C2 (µF)
RC/R3 (kΩ)
CC/C3 (nF)
L1 (µH)
1.2 22 47 3.24 6.8 3.3 1.8 22 47 6.8 6.8 3.3 2.5 22 47 6.8 6.8 10 3.3 22 47 6.8 6.8 10 5 22 47 6.8 6.8 10 12 22 47 6.8 6.8 15
Table 2 – Recommended Component Selection
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Application Information (cont.)
Inductor
Calculating the inductor value is a critical factor in designing a buck converter. For most designs, the following equation can be used to calculate the inductor value;
L=
VOUT⋅(VIN−VOUT)
VIN⋅ΔIL⋅fSW
WhereΔILis the inductor ripple current.
And fSW is the buck converter switching frequency.
Choose the inductor ripple current to be 30% of the maximum load current. The maximum inductor peak current is calculated from:
ΔI
IL(MAX)=ILOAD+L
2
Peak current determines the required saturation current rating, which influences the size of the inductor. Saturating the inductor decreases the converter efficiency while increasing the temperatures of the inductor and the internal MOSFETs. Hence choosing an inductor with appropriate saturation current rating is important.
A 1µH to 10µH inductor with a DC current rating of at least 25% percent higher than the maximum load current is recommended for most applications.
For highest efficiency, the inductor’s DC resistance should be less than 200mΩ. Use a larger inductance for improved efficiency under light load conditions.
Input Capacitor
The input capacitor reduces the surge current drawn from the input supply and the switching noise from the device. The input capacitor has to sustain the ripple current produced during the on time on the upper MOSFET. It must hence have a low ESR to minimize the losses.
The RMS current rating of the input capacitor is a critical parameter that must be higher than the RMS input current. As a rule of thumb, select an input capacitor which has an RMs rating that is greater than half of the maximum load current.
Due to large dI/dt through the input capacitors, electrolytic or ceramics should be used. If a tantalum must be used, it must be surge protected. Otherwise, capacitor failure could occur. For most applications, a 4.7µF ceramic capacitor is sufficient.
Output Capacitor
The output capacitor keeps the output voltage ripple small, ensures feedback loop stability and reduces the overshoot of the output voltage. The output capacitor is a basic component for the fast response of the power supply. In fact, during load transient, for the first few microseconds it supplies the current to the load. The converter recognizes the load transient and sets the duty cycle to maximum, but the current slope is limited by the inductor value.
Maximum capacitance required can be calculated from the following equation:
ESR of the output capacitor dominates the output voltage ripple. The amount of ripple can be calculated from the equation below:
Voutcapacitor=ΔIinductor*ESR
An output capacitor with ample capacitance and low ESR is the best option. For most applications, a 22µF ceramic capacitor will be sufficient.
ΔIinductor2
)2 Co=
22(Δ V+Vout)−Vout
L(Iout+
WhereΔVis the maximum output voltage overshoot.
AP6503
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Application Information (cont.)
PC Board Layout
This is a high switching frequency converter. Hence attention must be paid to the switching currents interference in the layout. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces.
External feedback resistor dividers must be placed close to the FB pin. 34mm
Input capacitor C1 must be placed as close as possible to the IC and to L1.
52mm
AP6503 is exposed at the bottom of the package and must be soldered directly to a well designed thermal pad on the PCB. This will help to increase the power dissipation.
External Bootstrap Diode
It is recommended that an external bootstrap diode be added when the input voltage is no greater than 5V or the 5V rail is available in the system. This helps to improve the efficiency of the regulator. This solution is also applicable for D > 65%. The bootstrap diode can be a low cost one such as BAT54 or a schottky that has a low Vf.
5VBSTAP6503SW34BOOST DIODE10nF
Figure 7 – External Bootstrap Compensation Components
Recommended Diodes:
Part Number
B130 SK13
Voltage/Current
Rating 30V, 1A 30V, 1A
Vendor Diodes Inc Diodes Inc
AP6503
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AP6503
Ordering Information
AP6503 SP - 13PackageSP : SO-8EP
Part Number AP6503SP-13
Package Code SP
Packaging SO-8EP
Packing13 : Tape & Reel
13” Tape and Reel
Quantity 2500/Tape & Reel
Part Number Suffix
-13
Marking Information
Package Outline Dimensions (All dimensions in mm.)
SO-8EP
85E114 Exposed PadHFbBottom ViewN7°A4° ± 3°9° (All sides)E45°QCGauge PlaneSeating PlaneeDA1E0LSO-8EP (SOP-8L-EP)Dim Min Max TypA 1.40 1.50 1.45A1 0.00 0.13 - b 0.30 0.50 0.40C 0.15 0.25 0.20D 4.85 4.95 4.90E 3.80 3.90 3.85E0 3.85 3.95 3.90E1 5.90 6.10 6.00e - - 1.27F 2.75 3.35 3.05H 2.11 2.71 2.41L 0.62 0.82 0.72N - - 0.35Q 0.60 0.70 0.65All Dimensions in mm
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Suggested Pad Layout
SO-8EP
X2
Y2Y1 X1 Y CX
AP6503
Document number: DS35077 Rev. 5 - 2
Dimensions
Value (in mm) C 1.270 X 0.802 X1 3.502 X2 4.612 Y 1.505 Y1 2.613 Y2
6.500
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